Microwave and Millimeter Wave Technologies: Modern UWB antennas and equipment doc
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I
Microwave and Millimeter Wave
Technologies: Modern UWB
antennas and equipment
Microwave and Millimeter Wave
Technologies: Modern UWB
antennas and equipment
Edited by
Prof. Igor Minin
In-Tech
intechweb.org
Published by In-Teh
In-Teh
Olajnica 19/2, 32000 Vukovar, Croatia
Abstracting and non-prot use of the material is permitted with credit to the source. Statements and
opinions expressed in the chapters are these of the individual contributors and not necessarily those of
the editors or publisher. No responsibility is accepted for the accuracy of information contained in the
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publication of which they are an author or editor, and the make other personal use of the work.
© 2010 In-teh
www.intechweb.org
Additional copies can be obtained from:
First published March 2010
Printed in India
Technical Editor: Sonja Mujacic
Cover designed by Dino Smrekar
Microwave and Millimeter Wave Technologies:
Modern UWB antennas and equipment,
Edited by Prof. Igor Minin
p. cm.
ISBN 978-953-7619-67-1
V
Preface
Novel generation wireless system is a packet switched wireless system with wide area
coverage and high throughput. It is designed to be cost effective and to provide high spectral
efciency . The 4th wireless uses Orthogonal Frequency Division Multiplexing, Ultra Wide
Radio Band and millimeter wireless and smart antenna. Highly directive, planar UWB
antennas are gaining more and more attention, as required in many novel and important
applications. With the release of the microwave and millimeter band, applications for short-
range and high-bandwidth different new devices are primary research areas in UWB systems.
Two different techniques for Directive Ultra-Wideband Planar Antennas were presented
throughout chapter 1: the operation of a novel bow-tie antenna with high front-to-back ratio
and directivity and a differential planar UWB antenna characterized by higher gain (more than
11 dB around 7 GHz) with respect to conventional printed radiators has been demonstrated.
The chapter 2 is timely in reporting the aspects of the conventional and state-of-the-art
antenna design in the UWB system. For example, design methods of the conventional UWB
antennas such as log-periodic dipole array are overviewed and the UWB antennas which
are recently researched are introduced. Also the principle and design methods to notch the
particular frequency band in UWB antenna are summarized
As it well known microstrip patch antennas have problems of low bandwidths. In the chapter
3 various ways to overcome this problem by using various matching techniques for numerous
patch antenna array schemes is show.
The recent progress in the development of UWB planar antenna technology has been reviewed
and some types of UWB metal-plate monopole antennas, UWB printed monopole antennas
and UWB printed slot antennas are described in the chapter 4.
In the chapter 5 design and implementation of a practical recongurable communication
system including an additive ultra wide band white Gaussian noise and delay lines in X-band
from 6 to 12 GHz with other necessary microwave parts as the test bed are introduced.
Also design and implementation procedures of all microwave parts such as ultra wideband
ampliers, dividers, switches, drivers, gain controllers, generators, lters, delay components,
bias tee, transitions and etcetera are discussed.
Slot array antennas using rectangular waveguides were widely used based on their various
important capabilities in microwave telecommunications. In the chapter 6 detailed study of
polarization agility achievement in Annular Waveguide Slot Antennas (AWSA) are presented.
In an AWSA, the waveguide and the slots are all circularly oriented to t the boundary.
VI
Far-eld radiation pattern control has strong potential in smart antennas, wireless
communications and radar. Typical planned applications include multipath fading and
interference mitigation, data rate and coverage enhancement, etc. For implementing these
functionalities, either switch beam or recongurable Half-Power Beam-Width (HPBW)
antennas are required. In the chapter 7, the authors present different antenna concepts to
obtain recongurable radiation pattern capability in millimeter waves as follows: multibeam
antennas are demonstrated based on Butler matrices at 24 and 60 GHz, and recongurable
Half Power BeamWidth antennas are shown. Also a new technique is presented to shape
the radiation pattern of an antenna to achieve directive or sectorial beams. In this case, the
antenna design is based on an inhomogeneous lens fed by several sources.
In the chapter 8 author investigate the ring loop antenna for the UHF digital terrestrial
broadcasting. A ring loop antenna is excited by a simple and low-cost feeder system. The
broadband input impedance and the high gain are obtained in the calculation and the
measurement.
During the past few decades, there has been growing interest in the use of microwave and
millimeter wave radiometers for remote sensing of the earth. The chapter 9 addresses the
subject of antenna array design in aperture synthesis radiometers. Also a novel antenna array
is described, which is a sparse antenna array with an offset parabolic cylinder reector at
millimeter wave band.
As it well known the antenna is an important element of communication, remote sensing and
radio-localization systems. The measurement of the antenna radiation pattern characteristics
allows to verify the conformity of the antenna. The different antenna measurement techniques
are reviewed in the chapter 10.
The interpretation of the radiated EMI measurement is a very complex problem due to many
disturbing inuences affecting such a measurement. The antenna brings into measurement
additional errors, which increase measurement uncertainty. These errors and their effect
on the entire uncertainty of the measurement are investigated in the chapter 11 in case of
broadband Bilog antenna, a typical receiving antenna for radiated EMI measurement.
Over the past few years, a large number of pattern synthesis techniques of antenna arrays
have been studied and developed. In the chapter 12 authors study the inuence of the sensor
directivity into the array beampattern, in order to test if the effects on the array pattern must
be taken into consideration in design methods of pattern array synthesis, and other array
design methods.
In the chapter 13, a brief introduction of mm-wave RoF system is given and the optical
techniques of generating mm-wave signals are presented. Unlike the conventional discussions
about mm-wave RoF systems focusing on the downlink only, the design of bidirectional mm-
wave RoF systems are also considered.
Rain-induced attenuation creates one of the most damaging effects of the atmosphere on
the quality of radio communication systems, especially those operating above 10 GHz. The
chapter 14 presents results that have thus far been acquired from an integrated research
campaign jointly carried out by researchers at Institut Teknologi Sepuluh Nopember,
Indonesia and Kumamoto University, Japan. The research is aimed at devising transmission
strategies suitable for broadband wireless access in microwave and millimeter-wave bands,
especially in tropical regions.
VII
High-precision and high-temporal global rainfall maps are very important for scientic
studies for global water cycle and practical applications for water resources. The purpose of
the chapter 15 is to briey describe the Global Satellite Mapping of Precipitation algorithm,
provide examples of rainfall maps from microwave radiometers.
Conventional metallic waveguides have several major advantages, including low propagation
losses and high power transmissions in the microwave frequency range. One disadvantage
of metallic waveguides is that the propagation frequency band is limited at frequencies
above the cutoff frequency. The chapter 16 introduces a system that uses a dual-frequency
band waveguide. Authors present the fundamental principles of this dual-frequency
band waveguide in which a dual in-line dielectric array is installed. It has been shown the
electromagnetic waves were propagated in a waveguide with dual in-line dielectric rods
made of LaAlO3 and without higher modes above the 2 cutoff frequencies.
In the chapter 17 authors propose a design method of a voltage-controlled oscillator using
on-chip coplanar waveguide (CPW) resonator thus replacing an LC-resonator at 5 GHz band.
It has been shown the advantages of employing CPW resonator is the wide frequency-tuning
range, and it also saves about 30% of chip size whereas the measured other performance of
the proposed oscillators are comparable to that of an oscillator using LC resonator. The design
technique is applicable for higher frequencies.
The microwave and millimeter wave broadband amplier is importance for wideband
wireless communications operating within microwave frequency range. The chapter 18
provides the fundamental design concepts of broadband amplier using the modern CMOS
technology. Various design techniques are introduced for achieving high performance
microwave broadband ampliers. The main design considerations and current trends are also
discussed.
The chapter 19 illustrates the interferometric concept in quadrature down-conversion for
communication and radar sensor applications.
In the chapter 20 describe a new method for the analysis of passive waveguide components,
composed of the cascade connection of planar junctions. This new method extracts the main
computations out of the frequency loop, thus reducing the overall CPU effort for solving the
frequency-domain problem.
The fundamental theory permitting the synthesis of the negative group delay cell is described
in details in the chapter 21. A time domain study based on a Gaussian wave pulse response,
the physical meaning of this phenomenon at microwave wavelengths is also provided and
a new concept of frequency-independent active phase shifter used in recent applications are
described.
In the chapter 22, the authors summarize the design procedure of broadband MMIC high
power ampliers. Some special considerations, as well as, experimental results are focused
on GaN technology.
Historically magnetrons were one of the rst devices used to build radar systems. In the
last chapter 23 it has been shown that the utilization of recent advances in magnetron
manufacturing technology, the introduction of novel approaches in radar design as well as a
vast progress in digital signal processing technique result in a solid overall performance of the
magnetron based millimeter wavelengths radars.
VIII
It is expected the book will attract more interest in microwave and millimeter wave
technologies and simulate new ideas on this fascinating subject.
Editor:
Prof. Igor Minin
Novosibirsk State Technical University, Russia
IX
Contents
Preface V
1. DirectiveUltra-WidebandPlanarAntennas 001
A D.Capobianco,F.M.Pigozzo,A.Locatelli,D.Modotto,C.DeAngelis,
S.Boscolo,F.Sacchetto,M.Midrio
2. Ultra-WidebandAntenna 019
CheolbokKim
3. PatchAntennasandMicrostripLines 049
JohnR.OjhaandMarcPeters
4. UWBandSWBPlanarAntennaTechnology 063
Shun-ShiZhong
5. ACompletePracticalUltrawidebandTestBedinX-Band 083
GholamrezaAskari,KhaterehGhasemiandHamidMirmohammadSadeghi
6. NovelPolarization-AgileAnnularWaveguideSlotAntennas 109
SiamakEbadiandKeyvanForooraghi
7. Recongurableradiationpatternantennasinmm-waves 123
OlivierLafond,M.Caillet,B.FuchsandM.Himdi
8. CharacteristicsofHigh-GainWidebandRingLoopAntennaanditsApplication 145
HaruoKawakami
9. AntennaArrayDesigninApertureSynthesisRadiometers 169
JianDongandQingxiaLi
10. AntennaMeasurement 193
DominiquePicard
11. Theinterferencebetweengroundplaneandreceivingantennaandits
effectontheradiatedEMImeasurementuncertainty 215
MikulasBittera,ViktorSmiesko,KarolKovacandJozefHallon
12. Analysisofdirectivesensorinuenceonarraybeampatterns 229
LaradelVal,AlbertoIzquierdo,MaríaI.Jiménez,JuanJ.VillacortaandMarianoRaboso
X
13. Millimeter-waveRadiooverFiberSystemforBroadband
WirelessCommunication 243
HaoshuoChen,RujianLinandJiajunYe
14. Measurementandmodelingofrainintensityandattenuationforthedesignand
evaluationofmicrowaveandmillimeter-wavecommunicationsystems 271
GamantyoHendrantoroandAkiraMatsushima
15. High-TemporalGlobalRainfallMapsfromSatellitePassive
MicrowaveRadiometers 301
ShoichiShige,SatoshiKida,TomoyaYamamoto,TakujiKubotaandKazumasaAonashi
16. ADual-FrequencyMetallicWaveguideSystem 313
YoshihiroKokubo
17. ApplicationsofOn-ChipCoplanarWaveguidestoDesignLocalOscillators
forWirelessCommunicationsSystem 329
RameshK.Pokharel,HaruichiKanayaandKeijiYoshida
18. DesignTechniquesforMicrowaveandMillimeterWaveCMOS
BroadbandAmpliers 345
ShawnS.H.HsuandJun-DeJin
19. MULTI-PORTTECHNOLOGYANDAPPLICATIONS 363
MoldovanEmilia,BosisioG.Renato,WuKeandTatuSeriojaO.
20. WidebandRepresentationofPassiveComponentsbasedonPlanarWaveguide
Junctions 389
FermínMira,ÁngelA.SanBlas,VicenteE.BoriaandBenitoGimeno
21. StudyandApplicationofMicrowaveActiveCircuitswithNegativeGroupDelay 415
BlaiseRavelo,AndréPérennecandMarcLeRoy
22. BroadbandGaNMMICPowerAmpliersdesign 441
María-ÁngelesGonzález-GarridoandJesúsGrajal
23. MagnetronBasedRadarSystemsforMillimeterWavelengthBand–Modern
ApproachesandProspects 459
VadymVolkov
DirectiveUltra-WidebandPlanarAntennas 1
DirectiveUltra-WidebandPlanarAntennas
A D.Capobianco,F.M.Pigozzo,A.Locatelli,D.Modotto,C.DeAngelis,S.Boscolo,F.
Sacchetto,M.Midrio
x
Directive Ultra-Wideband Planar Antennas
A D. Capobianco
#1
, F.M. Pigozzo
#2
, A. Locatelli
*3
, D. Modotto
*4
, C. De
Angelis
*5
, S. Boscolo
+6
, F. Sacchetto
+7
, M. Midrio
+8
#DEI, Università degli Studi di Padova
*DEA, Università degli Studi di Brescia
+DIEGM, Università degli Studi di Udine
Italy
1. Introduction
Since the acceptance of unlicensed use of the Ultra-Wideband (UWB) technology in the
range between 3.1 and 10.6 GHz in the USA (FCC, 2002) and more recently between 3.4 and
8.5 GHz in Europe (ETSI, 2008), the realization of low-cost UWB wireless systems is
considered a fundamental research goal both for military and commercial applications. The
possible use and benefits of UWB technology are significant and among its potential
applications, high-resolution radar and short-range ultra-high speed data transmission are
very attractive. In this scenario, design, fabrication and characterization of effective antennas
for UWB systems are challenging tasks with respect to the case of narrowband systems. A
suitable UWB antenna should be capable of operating over an ultra-wide bandwidth.
Therefore it is necessary to guarantee a good behavior of the antenna in the band of interest
in terms of impedance matching with the transmitter, radiation and time-domain properties.
Moreover, recent UWB antenna development tends to focus on ultra-compact planar
antennas as they are more practical in terms of manufacturing, integration with the system
electronics board and form factor. Typical configurations exhibit radiation patterns similar
to the traditional monopole/dipole antennas, i.e. they behave as omnidirectional radiators
in the plane normal to the radiating element. This feature is desirable in UWB devices which
do not have a fixed or a-priori defined orientation with respect to the environment and thus
when it is not necessary to favour any specific direction. On the other hand, strongly
directive radiators are required for radar applications, especially when low-power, low-
interference and high-resolution devices are needed. Directive UWB radiators are also
interesting towards several complementary goals, e.g. to provide extra radio link gain to
single antenna transceivers, to mitigate the effects of multipath in the indoor UWB channel,
and, last but not least, to result in a high front-to-back ratio, which is desirable in many
applications such as in wireless body-area networks (WBAN).
In the past few years, several printed broadband monopole-like configurations have been
reported for UWB applications, but presently, very few efforts have been made to increase
their directionality. This chapter intends to provide the reader with two different design
methodologies for increasing the directivity of planar UWB antennas. In section 2, a novel
antenna layout will be presented, as the result of subsequent modifications of a native
1
MicrowaveandMillimeterWaveTechnologies:ModernUWBantennasandequipment2
omnidirectional radiator: the bow-tie antenna. A high front-to-back ratio, low-profile design
will be developed by exploiting a planar reflector and studying ad-hoc optimizations of both
the antenna geometry and the feeding arrangement. The final layout could be particularly
well-suited for point-to-point high data-rate UWB radio links around the 5.5 GHz center
frequency.
In section 3, a different approach will be adopted: instead of backing an omnidirectional
printed wideband dipole, a structured ground plane will be employed with a two-element
array of disc monopoles. The resulting highly directive UWB antenna is designed to be used
in combination with a single-chip radar transceiver operating between 6 and 8 GHz
(Cacciatori et al., 2007).
The reader will find a detailed step-by-step design procedure, along with experimental data
obtained through characterization of several prototypes in anechoic chamber.
2. An UWB Bow-Tie Antenna with High Front-to-Back Ratio
2.1 Design Principles
As previously said in the introduction of this chapter, in the UWB communication
framework there are scenarios in which directivity is mandatory, for example when the
antenna has to be located in the corner of a room or against a wall to provide a sectoral radio
coverage from the transmitter (i.e. an access point) to the receiver (i.e. a set-top box).
Moreover, small dimensions and low-profile are desirable features for an easier integration
in the final device. For all these reasons the RF engineer may start the antenna design with
the choice of the candidate wideband radiator along with its initial geometrical parameters.
As directionality can be effectively achieved through the use of a planar reflector, the wide-
band, planar radiating element can be natively omnidirectional, such as the bow-tie antenna.
In the following, a design methodology is presented, starting from the technical
specifications summarized in Table 1.
center frequency 5.5 GHz
fractional bandwidth ≥ 20%
laminate Rogers RO4003C
laminate dimensions 30 x 50 mm
2
flat reflector dimensions 100 x 70 mm
2
laminate – reflector distance roughly 5 mm
Table 1. Technical specifications.
The well known design guidelines for a bow-tie resonating at 5.5 GHz (Balanis, 2005) would
lead to a total length of 16 mm, a small value that may cause problems when connecting the
antenna to the feed-line whose dimensions are comparable with the radiator itself; this fact
imposes to enlarge the antenna, and thus to force it to operate at a higher order resonance.
However, the laminate technical specifications limit the maximum size of the bow-tie. As a
consequence, a good trade-off is the antenna layout shown in Fig. 1. The latter will in turn
allow to save enough space to arrange the feeding/matching line on the same laminate. It is
important to note that an antipodal configuration is preferable since the antenna will be fed
using a 50 Ohm SMA connector.
Such an early model, mounted at a given distance above a planar reflector, can be studied
and subsequently optimized through numerical simulations with CST Microwave Studio
commercial software (CST, 2009).
Fig. 1. The bow-tie planar wideband dipole. The reflector is omitted for sake of clarity. The
dark (light) gray region is on the bottom (top) of the laminate, in antipodal configuration.
As shown in Fig. 2, the computed input impedance reveals a narrowband behavior, as the
reactance is nearly zero over a small frequency range around the highest resonance;
however, the real part is nearly constant between 6 and 8 GHz. This suggests to investigate a
modified design to move down the highest resonance, as well as the flat resistance region,
towards the 5.5 GHz center frequency.
0
200
400
600
800
1000
-400
-200
0
200
400
600
2 3 4 5 6 7 8 9 10
Re{Z } []
Frequency [GHz]
in
Im{Z } []
in
Fig. 2. Input impedance of the initial antenna of Fig. 1; real part (solid line) and imaginary
part (dashed line).
Aiming to keep untouched the reference bow-tie dimensions, the effect of moving the
feeding point off the vertex and along the edge of the patches can be numerically explored
(see Fig. 3). In Figs. 4(a) and 4(b) the real and imaginary part of the input impedance are
plotted versus frequency, for different values of the distance between the feeding point and
the vertex. It is interesting to note that when this distance (“a”parameter in Fig. 3) ranges
between 5 and 6 mm, the flat region of the real part of the input impedance falls in the band
of interest, and the resonance approaches the desired center frequency of 5.5 GHz.
DirectiveUltra-WidebandPlanarAntennas 3
omnidirectional radiator: the bow-tie antenna. A high front-to-back ratio, low-profile design
will be developed by exploiting a planar reflector and studying ad-hoc optimizations of both
the antenna geometry and the feeding arrangement. The final layout could be particularly
well-suited for point-to-point high data-rate UWB radio links around the 5.5 GHz center
frequency.
In section 3, a different approach will be adopted: instead of backing an omnidirectional
printed wideband dipole, a structured ground plane will be employed with a two-element
array of disc monopoles. The resulting highly directive UWB antenna is designed to be used
in combination with a single-chip radar transceiver operating between 6 and 8 GHz
(Cacciatori et al., 2007).
The reader will find a detailed step-by-step design procedure, along with experimental data
obtained through characterization of several prototypes in anechoic chamber.
2. An UWB Bow-Tie Antenna with High Front-to-Back Ratio
2.1 Design Principles
As previously said in the introduction of this chapter, in the UWB communication
framework there are scenarios in which directivity is mandatory, for example when the
antenna has to be located in the corner of a room or against a wall to provide a sectoral radio
coverage from the transmitter (i.e. an access point) to the receiver (i.e. a set-top box).
Moreover, small dimensions and low-profile are desirable features for an easier integration
in the final device. For all these reasons the RF engineer may start the antenna design with
the choice of the candidate wideband radiator along with its initial geometrical parameters.
As directionality can be effectively achieved through the use of a planar reflector, the wide-
band, planar radiating element can be natively omnidirectional, such as the bow-tie antenna.
In the following, a design methodology is presented, starting from the technical
specifications summarized in Table 1.
center frequency 5.5 GHz
fractional bandwidth ≥ 20%
laminate Rogers RO4003C
laminate dimensions 30 x 50 mm
2
flat reflector dimensions 100 x 70 mm
2
laminate – reflector distance roughly 5 mm
Table 1. Technical specifications.
The well known design guidelines for a bow-tie resonating at 5.5 GHz (Balanis, 2005) would
lead to a total length of 16 mm, a small value that may cause problems when connecting the
antenna to the feed-line whose dimensions are comparable with the radiator itself; this fact
imposes to enlarge the antenna, and thus to force it to operate at a higher order resonance.
However, the laminate technical specifications limit the maximum size of the bow-tie. As a
consequence, a good trade-off is the antenna layout shown in Fig. 1. The latter will in turn
allow to save enough space to arrange the feeding/matching line on the same laminate. It is
important to note that an antipodal configuration is preferable since the antenna will be fed
using a 50 Ohm SMA connector.
Such an early model, mounted at a given distance above a planar reflector, can be studied
and subsequently optimized through numerical simulations with CST Microwave Studio
commercial software (CST, 2009).
Fig. 1. The bow-tie planar wideband dipole. The reflector is omitted for sake of clarity. The
dark (light) gray region is on the bottom (top) of the laminate, in antipodal configuration.
As shown in Fig. 2, the computed input impedance reveals a narrowband behavior, as the
reactance is nearly zero over a small frequency range around the highest resonance;
however, the real part is nearly constant between 6 and 8 GHz. This suggests to investigate a
modified design to move down the highest resonance, as well as the flat resistance region,
towards the 5.5 GHz center frequency.
0
200
400
600
800
1000
-400
-200
0
200
400
600
2 3 4 5 6 7 8 9 10
Re{Z } []
Frequency [GHz]
in
Im{Z } []
in
Fig. 2. Input impedance of the initial antenna of Fig. 1; real part (solid line) and imaginary
part (dashed line).
Aiming to keep untouched the reference bow-tie dimensions, the effect of moving the
feeding point off the vertex and along the edge of the patches can be numerically explored
(see Fig. 3). In Figs. 4(a) and 4(b) the real and imaginary part of the input impedance are
plotted versus frequency, for different values of the distance between the feeding point and
the vertex. It is interesting to note that when this distance (“a”parameter in Fig. 3) ranges
between 5 and 6 mm, the flat region of the real part of the input impedance falls in the band
of interest, and the resonance approaches the desired center frequency of 5.5 GHz.
MicrowaveandMillimeterWaveTechnologies:ModernUWBantennasandequipment4
Fig. 3. Unconventional bow-tie feeding.
0
200
400
600
800
1000
4 4.5 5 5.5 6 6.5 7
Re{Z } []
in
Frequency [GHz]
-400
-200
0
200
400
600
4 4.5 5 5.5 6 6.5 7
Im{Z } []
in
Frequency [GHz]
Fig. 4. (a) real part and (b) imaginary part of the input impedance versus frequency for
different positions of the feeding point: a = 0 mm (solid line), a = 5 mm (dashed line), a = 6
mm (dashed-dotted line), a= 7 mm (gray line).
The next step in the design procedure is to provide a feasible microstrip transmission line to
drive the signal up to the feeding points. A transmission line with 200 Ohm intrinsic
impedance, a value which roughly coincides with the real part of the antenna input
impedance, may be designed by suitably choosing the stripes widths and spacing (see Fig.
5(a)). Notice that, as in the case of many other planar transmission lines, the intrinsic
impedance decreases for increasing strip widths. This in turn suggests that matching to a
conventional 50 Ohm connector can be obtained by suitably tapering the strip widths.
Further refinements of the antenna resonance may be obtained by acting on the bow-tie tips,
i.e. where the most intense currents flow. For instance, we observed that if we truncate the
tip edges as shown in Fig. 5(b), a slight increase of the resonant frequency takes place (see
Fig. 6, where the reflection coefficient of the antenna with and without the tips is reported).
If the antenna with truncated tips is chosen, two ways of designing the transmission line
may be envisaged. Indeed, the stripes may be bent upward or downward from the feeding
points. The latter option (see Fig. 7) is preferable, as a longer transmission line will in turn
result in a smoother tapered transition to the 50 Ohm connector.
(
a
)
(
b
)
Fig. 5. Selected feeding point position and stripes arrangement: (a) with and (b) without
bow-tie tips.
-40
-35
-30
-25
-20
-15
-10
-5
0
4 4.5 5 5.5 6 6.5 7
Reflection Coefficient [dB]
Frequency [GHz]
Fig. 6. Reflection coefficient of the antenna with off-vertex feeding: solid line refers to design
of Fig. 5(a), dashed line refers to design of Fig. 5(b). In both cases, the reference impedance
was set to 200 Ohm.
Bending the stripes downward will also cause the transmission line to run through the bow-
tie truncated tips. As we will show in section 2.3, if the spacing between the line stripes and
the truncated tips is small enough, two radiating slots (Balanis, 2005) are created. This will
provide an additional resonance that may be tuned to further increase the overall antenna
bandwidth.
2.2 Antenna Layout
Fig. 8 shows the final antenna layout. The laminate is a 50 x 30 x 1.575 mm
3
Rogers RO4003C
mounted above a 110 x 90 mm
2
rectangular metal reflector (not depicted in the figure for
sake of clarity) at a distance of 6 mm.
The thickness of the copper is 35 µm everywhere. A linear tapering of the feed-line ending
with 3 mm width for the top layer and 8 mm width for the bottom layer provides the
transition to the 50 Ohm SMA connector.
(
a
)
(
b
)
DirectiveUltra-WidebandPlanarAntennas 5
Fig. 3. Unconventional bow-tie feeding.
0
200
400
600
800
1000
4 4.5 5 5.5 6 6.5 7
Re{Z } []
in
Frequency [GHz]
-400
-200
0
200
400
600
4 4.5 5 5.5 6 6.5 7
Im{Z } []
in
Frequency [GHz]
Fig. 4. (a) real part and (b) imaginary part of the input impedance versus frequency for
different positions of the feeding point: a = 0 mm (solid line), a = 5 mm (dashed line), a = 6
mm (dashed-dotted line), a= 7 mm (gray line).
The next step in the design procedure is to provide a feasible microstrip transmission line to
drive the signal up to the feeding points. A transmission line with 200 Ohm intrinsic
impedance, a value which roughly coincides with the real part of the antenna input
impedance, may be designed by suitably choosing the stripes widths and spacing (see Fig.
5(a)). Notice that, as in the case of many other planar transmission lines, the intrinsic
impedance decreases for increasing strip widths. This in turn suggests that matching to a
conventional 50 Ohm connector can be obtained by suitably tapering the strip widths.
Further refinements of the antenna resonance may be obtained by acting on the bow-tie tips,
i.e. where the most intense currents flow. For instance, we observed that if we truncate the
tip edges as shown in Fig. 5(b), a slight increase of the resonant frequency takes place (see
Fig. 6, where the reflection coefficient of the antenna with and without the tips is reported).
If the antenna with truncated tips is chosen, two ways of designing the transmission line
may be envisaged. Indeed, the stripes may be bent upward or downward from the feeding
points. The latter option (see Fig. 7) is preferable, as a longer transmission line will in turn
result in a smoother tapered transition to the 50 Ohm connector.
(
a
)
(
b
)
Fig. 5. Selected feeding point position and stripes arrangement: (a) with and (b) without
bow-tie tips.
-40
-35
-30
-25
-20
-15
-10
-5
0
4 4.5 5 5.5 6 6.5 7
Reflection Coefficient [dB]
Frequency [GHz]
Fig. 6. Reflection coefficient of the antenna with off-vertex feeding: solid line refers to design
of Fig. 5(a), dashed line refers to design of Fig. 5(b). In both cases, the reference impedance
was set to 200 Ohm.
Bending the stripes downward will also cause the transmission line to run through the bow-
tie truncated tips. As we will show in section 2.3, if the spacing between the line stripes and
the truncated tips is small enough, two radiating slots (Balanis, 2005) are created. This will
provide an additional resonance that may be tuned to further increase the overall antenna
bandwidth.
2.2 Antenna Layout
Fig. 8 shows the final antenna layout. The laminate is a 50 x 30 x 1.575 mm
3
Rogers RO4003C
mounted above a 110 x 90 mm
2
rectangular metal reflector (not depicted in the figure for
sake of clarity) at a distance of 6 mm.
The thickness of the copper is 35 µm everywhere. A linear tapering of the feed-line ending
with 3 mm width for the top layer and 8 mm width for the bottom layer provides the
transition to the 50 Ohm SMA connector.
(
a
)
(
b
)
MicrowaveandMillimeterWaveTechnologies:ModernUWBantennasandequipment6
Fig. 7. Schematic of the candidate design with off-vertex feeding point, truncated bow-tie
tips and bent feed-lines.
Fig. 8. Model and dimensions (in millimeters) of the proposed antenna. The left half of the
planar radiator is on the top of the substrate, whereas the right half is on the bottom
(antipodal configuration).
The parallel stripes which run between the bow-tie tips have a width equal to 0.6 mm. They
are printed on opposite sides of the substrate and are separated by a 0.3 mm gap (with
respect to the x-axis). The stripes are bent at a right angle and connect to the triangular
patches via two 7.4 mm-long branches. The central vertex of each triangular patch is “cut
away”, resulting in a 0.5 x 1.8 mm
2
slot between the feed-line and the patch. The entire
structure has been numerically modeled with CST Microwave Studio. All the elements have
been included in the simulations: the metallic patches, stripes and planar reflector, as well as
the SMA connector and the ending part of a 50 Ohm
coaxial cable filled with lossy Teflon.
All the metal parts have been regarded as perfect electric conductors, while the dielectrics in
the laminate and in the coaxial cable have been modeled as lossy materials. Values
r
= 3.38
and
r
= 2.1 have been used for the relative permittivity of RO4003C and Teflon. As for the
losses,
tan = 0.0027 and tan = 0.001 have been used, respectively. Simulations have been
performed in the 4 to 7 GHz range in order to optimize all the geometrical parameters of the
antenna, including the dimensions of the stripes, the transition and the distance between the
laminate and the reflector.
With the dimensions reported in Fig. 8, a prototype (Fig. 9) was fabricated using a
numerically-controlled milling machine. Reflection coefficient and radiation patterns of the
antenna under test have been measured in anechoic chamber using a 2-port Agilent N5230A
PNA-L network analyzer, a Satimo SGH-820 ridged horn wideband probe antenna, and a
remotely controlled turntable.
Fig. 9. Photograph of the antenna that has been fabricated and experimentally characterized.
2.3 The Radiation Mechanism
Figs. 10 and 11 show the simulated (dashed-dotted line) and measured (solid line) reflection
coefficient (S
11
parameter) and its representation on the Smith chart for the proposed
antenna, as normalized to 50 Ohm. The measured band at |S
11
|= -10 dB spans from 4.8 GHz
to 6.1 GHz. Therefore the bandwidth equals about a fourth of the center frequency, thus
making the proposed bow-tie an Ultra-Wideband antenna, according to the FCC definition
(FCC, 2002).
Fig. 10. Simulated (solid line) and measured (dashed-dotted line) magnitude of the
reflection coefficient (|S
11
|) versus frequency.
As for the radiation mechanism, we notice that the simulations clearly show two separate
resonances which concur to form the antenna overall bandwidth. At the lower resonance, an
DirectiveUltra-WidebandPlanarAntennas 7
Fig. 7. Schematic of the candidate design with off-vertex feeding point, truncated bow-tie
tips and bent feed-lines.
Fig. 8. Model and dimensions (in millimeters) of the proposed antenna. The left half of the
planar radiator is on the top of the substrate, whereas the right half is on the bottom
(antipodal configuration).
The parallel stripes which run between the bow-tie tips have a width equal to 0.6 mm. They
are printed on opposite sides of the substrate and are separated by a 0.3 mm gap (with
respect to the x-axis). The stripes are bent at a right angle and connect to the triangular
patches via two 7.4 mm-long branches. The central vertex of each triangular patch is “cut
away”, resulting in a 0.5 x 1.8 mm
2
slot between the feed-line and the patch. The entire
structure has been numerically modeled with CST Microwave Studio. All the elements have
been included in the simulations: the metallic patches, stripes and planar reflector, as well as
the SMA connector and the ending part of a 50 Ohm
coaxial cable filled with lossy Teflon.
All the metal parts have been regarded as perfect electric conductors, while the dielectrics in
the laminate and in the coaxial cable have been modeled as lossy materials. Values
r
= 3.38
and
r
= 2.1 have been used for the relative permittivity of RO4003C and Teflon. As for the
losses,
tan = 0.0027 and tan = 0.001 have been used, respectively. Simulations have been
performed in the 4 to 7 GHz range in order to optimize all the geometrical parameters of the
antenna, including the dimensions of the stripes, the transition and the distance between the
laminate and the reflector.
With the dimensions reported in Fig. 8, a prototype (Fig. 9) was fabricated using a
numerically-controlled milling machine. Reflection coefficient and radiation patterns of the
antenna under test have been measured in anechoic chamber using a 2-port Agilent N5230A
PNA-L network analyzer, a Satimo SGH-820 ridged horn wideband probe antenna, and a
remotely controlled turntable.
Fig. 9. Photograph of the antenna that has been fabricated and experimentally characterized.
2.3 The Radiation Mechanism
Figs. 10 and 11 show the simulated (dashed-dotted line) and measured (solid line) reflection
coefficient (S
11
parameter) and its representation on the Smith chart for the proposed
antenna, as normalized to 50 Ohm. The measured band at |S
11
|= -10 dB spans from 4.8 GHz
to 6.1 GHz. Therefore the bandwidth equals about a fourth of the center frequency, thus
making the proposed bow-tie an Ultra-Wideband antenna, according to the FCC definition
(FCC, 2002).
Fig. 10. Simulated (solid line) and measured (dashed-dotted line) magnitude of the
reflection coefficient (|S
11
|) versus frequency.
As for the radiation mechanism, we notice that the simulations clearly show two separate
resonances which concur to form the antenna overall bandwidth. At the lower resonance, an
MicrowaveandMillimeterWaveTechnologies:ModernUWBantennasandequipment8
intense electric field is observed between the parallel stripes which run in the middle of the
bow-tie tips. The stripes form a slot, loaded by the metallic patches which are connected at
the slot far end by the T-branches. The resonant frequency scales inversely with the length
of the parallel stripes.
Fig. 11. Input impedance of the antenna on the Smith chart. Dashed-dotted line and solid
line are used for simulated and measured data, respectively.
At the upper resonant frequency, radiation is due to the electric field which seeds in the
slots etched between the parallel stripes and the bow-tie tips. The resonant frequency is
influenced by the length of both the parallel stripes, and of the T-branches.
Fig. 12. Simulated surface magnetic currents intensity at the lower resonant frequency.
Figs. 12 and 13 confirm this picture by showing the numerically evaluated electric field at
the two resonances. At the lower resonance (Fig. 12), a large electric field is observed all
along the right-hand side of the feeding strip. Whereas, at the upper resonance (Fig. 13), it is
clearly visible that the field no longer reaches the T-branch at the end of the strip. It is much
more tightly confined in the region between the left-hand side of the strip and the truncated
bow-tie tip, and rapidly vanishes away from the slot.
Fig. 13. Simulated surface magnetic currents intensity at the upper resonant frequency.
In the proposed antenna, therefore, the bow-tie is not used to realize a wideband radiator, as
it is normally done in “conventional” bow-tie dipoles (Eldek et al., 2005), (Kiminami et al.,
2004). Rather, the metallic patches are used as wideband loading or matching networks, and
radiation is primarily due to magnetic currents which flow in properly designed slots.
Notice also that the two separate resonances we have observed in the simulations do not
appear in the experimental results as clearly as in the simulations. We explain this
discrepancy on the basis of the following considerations. When we numerically simulated
the antenna, we observed that the spectral location of the upper resonant frequency is very
sensitive to the electrical length of the T-branches. That is, small variations of either the
physical length of these metallic stripes, or of the dielectric constant of the underlying slab
may cause a noticeable shift of the upper resonant frequency. If the frequency shifts
upwards, |S
11
| between the two resonances may eventually exceed the –10 dB level, finally
splitting the operation bandwidth into two separate regions. Whereas, if the frequency shifts
downward, the upper resonance tends to merge with the lower one, thus forming an unique
dip in the |S
11
| plot. These behaviours are accompanied by easily detectable features on the
Smith chart. Indeed, the two resonances form a loop near the chart center. The loop gets
larger when the resonant frequencies split. Viceversa, it tends to collapse and to eventually
disappear when the resonant frequencies merge. This is what happened in our experiment:
the loop in the vicinity of the chart center has a very reduced extension, revealing that the
two resonant frequencies have merged. The likely cause is either a misfabrication, or a slight
discrepancy between the experimental and declared values of the dielectric constant.
2.4 Radiation Patterns, Front-to-Back Ratio, and Group Delay
Figs. 14, 15 and 16 show the radiation patterns in the x-z and y-z planes for the 4.8 GHz, 5.5
GHz and 6.1 GHz frequencies respectively. The -3 dB beamwidth ranges between 52 and 87
degrees and between 60 and 72 degrees in the x-z and y-z planes respectively. It may be
observed that the direction of maximum radiation is slightly off the normal to the plane
which contains the antenna. This is due to the antipodal layout of the antenna itself.
Fig. 17 shows the measured frequency dependence of the front-to-back ratio, which always
exceeds the level of 24 dB. The measured antenna gain ranges between 8 and 10 dB.
As for the antenna performance in the time-domain, we calculated the variation in
frequency of the group delay, which is related to the first-derivative of the measured phase
of the S
12
parameter. As can be seen in Fig. 18 the variation of the group delay in the band of
DirectiveUltra-WidebandPlanarAntennas 9
intense electric field is observed between the parallel stripes which run in the middle of the
bow-tie tips. The stripes form a slot, loaded by the metallic patches which are connected at
the slot far end by the T-branches. The resonant frequency scales inversely with the length
of the parallel stripes.
Fig. 11. Input impedance of the antenna on the Smith chart. Dashed-dotted line and solid
line are used for simulated and measured data, respectively.
At the upper resonant frequency, radiation is due to the electric field which seeds in the
slots etched between the parallel stripes and the bow-tie tips. The resonant frequency is
influenced by the length of both the parallel stripes, and of the T-branches.
Fig. 12. Simulated surface magnetic currents intensity at the lower resonant frequency.
Figs. 12 and 13 confirm this picture by showing the numerically evaluated electric field at
the two resonances. At the lower resonance (Fig. 12), a large electric field is observed all
along the right-hand side of the feeding strip. Whereas, at the upper resonance (Fig. 13), it is
clearly visible that the field no longer reaches the T-branch at the end of the strip. It is much
more tightly confined in the region between the left-hand side of the strip and the truncated
bow-tie tip, and rapidly vanishes away from the slot.
Fig. 13. Simulated surface magnetic currents intensity at the upper resonant frequency.
In the proposed antenna, therefore, the bow-tie is not used to realize a wideband radiator, as
it is normally done in “conventional” bow-tie dipoles (Eldek et al., 2005), (Kiminami et al.,
2004). Rather, the metallic patches are used as wideband loading or matching networks, and
radiation is primarily due to magnetic currents which flow in properly designed slots.
Notice also that the two separate resonances we have observed in the simulations do not
appear in the experimental results as clearly as in the simulations. We explain this
discrepancy on the basis of the following considerations. When we numerically simulated
the antenna, we observed that the spectral location of the upper resonant frequency is very
sensitive to the electrical length of the T-branches. That is, small variations of either the
physical length of these metallic stripes, or of the dielectric constant of the underlying slab
may cause a noticeable shift of the upper resonant frequency. If the frequency shifts
upwards, |S
11
| between the two resonances may eventually exceed the –10 dB level, finally
splitting the operation bandwidth into two separate regions. Whereas, if the frequency shifts
downward, the upper resonance tends to merge with the lower one, thus forming an unique
dip in the |S
11
| plot. These behaviours are accompanied by easily detectable features on the
Smith chart. Indeed, the two resonances form a loop near the chart center. The loop gets
larger when the resonant frequencies split. Viceversa, it tends to collapse and to eventually
disappear when the resonant frequencies merge. This is what happened in our experiment:
the loop in the vicinity of the chart center has a very reduced extension, revealing that the
two resonant frequencies have merged. The likely cause is either a misfabrication, or a slight
discrepancy between the experimental and declared values of the dielectric constant.
2.4 Radiation Patterns, Front-to-Back Ratio, and Group Delay
Figs. 14, 15 and 16 show the radiation patterns in the x-z and y-z planes for the 4.8 GHz, 5.5
GHz and 6.1 GHz frequencies respectively. The -3 dB beamwidth ranges between 52 and 87
degrees and between 60 and 72 degrees in the x-z and y-z planes respectively. It may be
observed that the direction of maximum radiation is slightly off the normal to the plane
which contains the antenna. This is due to the antipodal layout of the antenna itself.
Fig. 17 shows the measured frequency dependence of the front-to-back ratio, which always
exceeds the level of 24 dB. The measured antenna gain ranges between 8 and 10 dB.
As for the antenna performance in the time-domain, we calculated the variation in
frequency of the group delay, which is related to the first-derivative of the measured phase
of the S
12
parameter. As can be seen in Fig. 18 the variation of the group delay in the band of
MicrowaveandMillimeterWaveTechnologies:ModernUWBantennasandequipment10
interest is less than 250 ps, which is appreciably low as compared to the duration of the
stimulus in a pulsed transmission (about 1.5 ns for the 4.8 - 6.1 GHz band).
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
x
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
y
Fig. 14. Measured radiation patterns at 4.8 GHz; (a) in the x-z plane, (b) in the y-z plane.
Gray line: in-plane electric field; dashed-line: electric field orthogonal to the plane; solid-
line: total electric field. The results are reported in dB.
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
x
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
y
Fig. 15. Measured radiation patterns at 5.5 GHz; (a) in the x-z plane, (b) in the y-z plane.
Gray line: in-plane electric field; dashed-line: electric field orthogonal to the plane; solid-
line: total electric field. The results are reported in dB.
(a) (b)
(
a
)
(
b
)
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
x
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
y
Fig. 16. Measured radiation patterns at 6.1 GHz; (a) in the x-z plane, (b) in the y-z plane.
Gray line: in-plane electric field; dashed-line: electric field orthogonal to the plane; solid-
line: total electric field. The results are reported in dB.
0
5
10
15
20
25
30
35
5 5.25 5.5 5.75 6
Front-to-back ratio [dB]
Frequency [GHz]
Fig. 17. Measured front-to-back ratio in the frequency range 4.8 – 6.1 GHz.
(
a
)
(
b
)
DirectiveUltra-WidebandPlanarAntennas 11
interest is less than 250 ps, which is appreciably low as compared to the duration of the
stimulus in a pulsed transmission (about 1.5 ns for the 4.8 - 6.1 GHz band).
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
x
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
y
Fig. 14. Measured radiation patterns at 4.8 GHz; (a) in the x-z plane, (b) in the y-z plane.
Gray line: in-plane electric field; dashed-line: electric field orthogonal to the plane; solid-
line: total electric field. The results are reported in dB.
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
x
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
y
Fig. 15. Measured radiation patterns at 5.5 GHz; (a) in the x-z plane, (b) in the y-z plane.
Gray line: in-plane electric field; dashed-line: electric field orthogonal to the plane; solid-
line: total electric field. The results are reported in dB.
(
a
)
(
b
)
(
a
)
(
b
)
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
x
-40
-30
-20
-10
0
10
0
30
60
90
120
150
180
210
240
330
z
y
Fig. 16. Measured radiation patterns at 6.1 GHz; (a) in the x-z plane, (b) in the y-z plane.
Gray line: in-plane electric field; dashed-line: electric field orthogonal to the plane; solid-
line: total electric field. The results are reported in dB.
0
5
10
15
20
25
30
35
5 5.25 5.5 5.75 6
Front-to-back ratio [dB]
Frequency [GHz]
Fig. 17. Measured front-to-back ratio in the frequency range 4.8 – 6.1 GHz.
(
a
)
(
b
)
MicrowaveandMillimeterWaveTechnologies:ModernUWBantennasandequipment12
-1
-0.5
0
0.5
1
5 5.25 5.5 5.75 6
Group Delay [ns]
Frequency [GHz]
Fig. 18. Measured group delay in the frequency range 4.8 – 6.1 GHz.
3. A Planar, Differential, and Directive Ultra-Wideband Antenna
In this section a planar differential UWB antenna characterized by high directivity is
described. The proposed structure is composed of two disc monopoles with L-shaped
ground planes, fed by 50 Ohm microstrip lines (100 Ohm differential input). The
introduction of a structured ground plane and the array effect permit to achieve a measured
gain exceeding 11 dB even on a low-cost substrate, maintaining at the same time a
reasonable size of the board. As reference application, the antenna has been optimized to
work between 6 and 8 GHz in combination with a single-chip radar transceiver (Cacciatori
et al., 2007).
3.1 Antenna Layout
A sketch of the differential UWB antenna is reported in Fig. 19. The structure is formed by
two disc monopoles with L-shaped ground planes, fed by 50 Ohm microstrip lines (width
equal to 3 mm), on a low-cost FR4 substrate. We assumed that the FR4 dielectric constant is
4.5, and the declared board thickness is equal to 1.6 mm. Optimization of the antenna was
performed through CST Microwave Studio simulations, and the reported results correspond
to the device with the geometric parameters that guarantee the best theoretical performance.
In Fig. 19 we show the size of the board, and the position of the centers of the two disc
monopoles, whose radius is 9 mm.
The basic element of the proposed differential antenna is the single disc monopole, with the
related 50 Ohm microstrip feed-line and the L-shaped ground plane with rounded corner.
The distance between the edge of each disc and the horizontal and vertical metal stripes that
compose the L-shaped ground plane is 1 and 2 mm respectively. This building block (50 x 50
mm
2
) is then rotated by 40 degrees in the x-y plane, and it is depicted within a yellow square
in Fig. 19. The second element of the array is obtained by mirroring the first one with respect
to the x = 0 plane. The region between the two elements on the ground side is filled with
metal, so that this large common ground plane can be exploited for a straightforward
placement of a single-chip transceiver and its circuitry. As shown in Fig. 19, the distance
between the two disc monopoles is about 70 mm, which is larger than the maximum signal
wavelength. In Fig. 20 a photograph of the fabricated antenna is reported.
Fig. 19. Sketch of the differential UWB antenna: disc monopoles and microstrip lines (black),
structured ground plane (dark gray) and the basic element of the array (yellow square) are
depicted. The dimensions are in millimeters.
3.2 Reflection Coefficient, Group Delay and Radiation Patterns
Differential and common-mode reflection coefficients (|S
dd
| and |S
cc
|) have been
measured by using a 4-port vector network analyzer Anritsu MS4624D, whereas the single-
port reflection coefficient at port 1 (|S
11
|) has been measured through a 2-port Agilent
N5230A PNA-L network analyzer. In Fig. 21 we report results obtained from measurements
between 4 and 9 GHz; it is worth noting that the three curves are almost overlapped and this
implies that, as desired, coupling between the two antenna elements is very weak. The
differential reflection coefficient is below -10 dB in the frequency range between 5.3 and 9
GHz, thus the differential UWB antenna exhibits a good behavior in terms of impedance
matching over a bandwidth larger than 4 GHz. Also the group delay has been measured in
anechoic chamber, by using the same setup we have used for measurements of gain and
radiation patterns (see below for the setup description). The results are reported in Fig. 22: it
is fundamental to note that the maximum group delay variation is equal to fractions of a
nanosecond, therefore the antenna is able to guarantee low pulse distortion.
The radiative properties of the differential UWB antenna have been characterized in
anechoic chamber by utilizing a 2-port Agilent N5230A PNA-L network analyzer, two 180-
degree hybrid couplers, a Satimo SGH-820 ridged horn wideband probe antenna, and a
remotely controlled turntable. Here we report the results for co-polarization at 7 GHz; we
emphasize the fact that the level of cross-polarization is quite low, and the shape of the
radiation patterns is rather uniform over the entire band.
DirectiveUltra-WidebandPlanarAntennas 13
-1
-0.5
0
0.5
1
5 5.25 5.5 5.75 6
Group Delay [ns]
Frequency [GHz]
Fig. 18. Measured group delay in the frequency range 4.8 – 6.1 GHz.
3. A Planar, Differential, and Directive Ultra-Wideband Antenna
In this section a planar differential UWB antenna characterized by high directivity is
described. The proposed structure is composed of two disc monopoles with L-shaped
ground planes, fed by 50 Ohm microstrip lines (100 Ohm differential input). The
introduction of a structured ground plane and the array effect permit to achieve a measured
gain exceeding 11 dB even on a low-cost substrate, maintaining at the same time a
reasonable size of the board. As reference application, the antenna has been optimized to
work between 6 and 8 GHz in combination with a single-chip radar transceiver (Cacciatori
et al., 2007).
3.1 Antenna Layout
A sketch of the differential UWB antenna is reported in Fig. 19. The structure is formed by
two disc monopoles with L-shaped ground planes, fed by 50 Ohm microstrip lines (width
equal to 3 mm), on a low-cost FR4 substrate. We assumed that the FR4 dielectric constant is
4.5, and the declared board thickness is equal to 1.6 mm. Optimization of the antenna was
performed through CST Microwave Studio simulations, and the reported results correspond
to the device with the geometric parameters that guarantee the best theoretical performance.
In Fig. 19 we show the size of the board, and the position of the centers of the two disc
monopoles, whose radius is 9 mm.
The basic element of the proposed differential antenna is the single disc monopole, with the
related 50 Ohm microstrip feed-line and the L-shaped ground plane with rounded corner.
The distance between the edge of each disc and the horizontal and vertical metal stripes that
compose the L-shaped ground plane is 1 and 2 mm respectively. This building block (50 x 50
mm
2
) is then rotated by 40 degrees in the x-y plane, and it is depicted within a yellow square
in Fig. 19. The second element of the array is obtained by mirroring the first one with respect
to the x = 0 plane. The region between the two elements on the ground side is filled with
metal, so that this large common ground plane can be exploited for a straightforward
placement of a single-chip transceiver and its circuitry. As shown in Fig. 19, the distance
between the two disc monopoles is about 70 mm, which is larger than the maximum signal
wavelength. In Fig. 20 a photograph of the fabricated antenna is reported.
Fig. 19. Sketch of the differential UWB antenna: disc monopoles and microstrip lines (black),
structured ground plane (dark gray) and the basic element of the array (yellow square) are
depicted. The dimensions are in millimeters.
3.2 Reflection Coefficient, Group Delay and Radiation Patterns
Differential and common-mode reflection coefficients (|S
dd
| and |S
cc
|) have been
measured by using a 4-port vector network analyzer Anritsu MS4624D, whereas the single-
port reflection coefficient at port 1 (|S
11
|) has been measured through a 2-port Agilent
N5230A PNA-L network analyzer. In Fig. 21 we report results obtained from measurements
between 4 and 9 GHz; it is worth noting that the three curves are almost overlapped and this
implies that, as desired, coupling between the two antenna elements is very weak. The
differential reflection coefficient is below -10 dB in the frequency range between 5.3 and 9
GHz, thus the differential UWB antenna exhibits a good behavior in terms of impedance
matching over a bandwidth larger than 4 GHz. Also the group delay has been measured in
anechoic chamber, by using the same setup we have used for measurements of gain and
radiation patterns (see below for the setup description). The results are reported in Fig. 22: it
is fundamental to note that the maximum group delay variation is equal to fractions of a
nanosecond, therefore the antenna is able to guarantee low pulse distortion.
The radiative properties of the differential UWB antenna have been characterized in
anechoic chamber by utilizing a 2-port Agilent N5230A PNA-L network analyzer, two 180-
degree hybrid couplers, a Satimo SGH-820 ridged horn wideband probe antenna, and a
remotely controlled turntable. Here we report the results for co-polarization at 7 GHz; we
emphasize the fact that the level of cross-polarization is quite low, and the shape of the
radiation patterns is rather uniform over the entire band.
MicrowaveandMillimeterWaveTechnologies:ModernUWBantennasandequipment14
Fig. 20. Photograph of the fabricated antenna in anechoic chamber.
In Fig. 23(a) we show measured and simulated radiation pattern in the x-y plane when only
port 1 is excited. The results show the typical behavior of a disc monopole with L-shaped
ground plane, with a single lobe of radiation oriented along the y-axis, and the maximum
gain is equal to 8.6 dB. In Figs. 23(b) and 24(a) we report measured and simulated radiation
patterns in the x-y and y-z plane respectively, when the two input ports are excited with
signals with opposite polarity. The increase of directionality is clearly visible even from a
simple analysis of the shape of the patterns: indeed, the maximum gain is equal to 11.6 dB.
In Fig. 24(b) we report, for the sake of comparison, measured and simulated radiation
patterns for in-phase excitation of the two ports in the x-y plane. It is possible to note the
birth of two lobes, with a maximum gain equal to 9.1 dB. In all the cases, it is worth noting
the excellent agreement between numerical and experimental data. These results clearly
demonstrate that the proposed differential UWB antenna is a good candidate to be used
whenever a highly directive differential radiator with planar profile is required.
3.3 The Radiation Mechanism
We have numerically investigated the pattern of the surface currents on the UWB
differential antenna, in order to identify the reasons for such a huge increase of
directionality. The usage of a disc monopole with L-shaped ground plane guarantees the
suppression of one lobe of radiation with respect to conventional monopoles, and this
corresponds to an increase of gain of about 3 dB. The two disc monopoles that compose the
differential antenna can be considered as an array of two elements, wherein each element
radiates with only one main lobe of radiation along the y-direction. The structure should
ideally behave as a broad-side array, with maximum value of the array factor in normal
direction with respect to the alignment direction (i.e. x-axis) (Someda, 1998). From the
theory of antenna arrays, it is well known that we get a broad-side array when the current
distribution on each antenna element is in phase.
-20
-15
-10
-5
0
4 5 6 7 8 9
Reflection Coefficient [dB]
Frequency [GHz]
Fig. 21. Measured differential (solid line) and common-mode (dashed-dotted line) reflection
coefficients |S
dd
| and |S
cc
|, and measured single-port reflection coefficient |S
11
| (dotted
line) between 4 and 9 GHz.
-1
-0.5
0
0.5
1
5.5 6 6.5 7 7.5 8 8.5
Group Delay [ns]
Frequency [GHz]
Fig. 22. Measured group delay curve between 5.5 and 8.5 GHz.
In Fig. 25 we plot the y-component of the surface currents at 7 GHz for differential input (the
two ports have opposite polarity). It is straightforward to see that the currents on the
microstrips have opposite polarity, whereas on the radiating elements they are
predominantly distributed on the edge of the discs and are in phase due to the mirror
symmetry between the two elements. As a consequence, in this case the differential antenna
behaves as a broad-side array, as shown by the radiation pattern in Fig. 23(b). Viceversa,
when the two input ports are fed in phase the currents on the two microstrips are obviously
in phase, whereas on the radiating elements they are out of phase. Indeed, the radiation
pattern in Fig. 24(b) exhibits two radiation lobes and no enhancement of directivity, as
predicted by the calculation of the array factor. Notice that the increase of gain in the case of
differential input with respect to excitation of a single port is about 3 dB, which is exactly
the value predicted by the theory of antenna arrays for a two-element array.
DirectiveUltra-WidebandPlanarAntennas 15
Fig. 20. Photograph of the fabricated antenna in anechoic chamber.
In Fig. 23(a) we show measured and simulated radiation pattern in the x-y plane when only
port 1 is excited. The results show the typical behavior of a disc monopole with L-shaped
ground plane, with a single lobe of radiation oriented along the y-axis, and the maximum
gain is equal to 8.6 dB. In Figs. 23(b) and 24(a) we report measured and simulated radiation
patterns in the x-y and y-z plane respectively, when the two input ports are excited with
signals with opposite polarity. The increase of directionality is clearly visible even from a
simple analysis of the shape of the patterns: indeed, the maximum gain is equal to 11.6 dB.
In Fig. 24(b) we report, for the sake of comparison, measured and simulated radiation
patterns for in-phase excitation of the two ports in the x-y plane. It is possible to note the
birth of two lobes, with a maximum gain equal to 9.1 dB. In all the cases, it is worth noting
the excellent agreement between numerical and experimental data. These results clearly
demonstrate that the proposed differential UWB antenna is a good candidate to be used
whenever a highly directive differential radiator with planar profile is required.
3.3 The Radiation Mechanism
We have numerically investigated the pattern of the surface currents on the UWB
differential antenna, in order to identify the reasons for such a huge increase of
directionality. The usage of a disc monopole with L-shaped ground plane guarantees the
suppression of one lobe of radiation with respect to conventional monopoles, and this
corresponds to an increase of gain of about 3 dB. The two disc monopoles that compose the
differential antenna can be considered as an array of two elements, wherein each element
radiates with only one main lobe of radiation along the y-direction. The structure should
ideally behave as a broad-side array, with maximum value of the array factor in normal
direction with respect to the alignment direction (i.e. x-axis) (Someda, 1998). From the
theory of antenna arrays, it is well known that we get a broad-side array when the current
distribution on each antenna element is in phase.
-20
-15
-10
-5
0
4 5 6 7 8 9
Reflection Coefficient [dB]
Frequency [GHz]
Fig. 21. Measured differential (solid line) and common-mode (dashed-dotted line) reflection
coefficients |S
dd
| and |S
cc
|, and measured single-port reflection coefficient |S
11
| (dotted
line) between 4 and 9 GHz.
-1
-0.5
0
0.5
1
5.5 6 6.5 7 7.5 8 8.5
Group Delay [ns]
Frequency [GHz]
Fig. 22. Measured group delay curve between 5.5 and 8.5 GHz.
In Fig. 25 we plot the y-component of the surface currents at 7 GHz for differential input (the
two ports have opposite polarity). It is straightforward to see that the currents on the
microstrips have opposite polarity, whereas on the radiating elements they are
predominantly distributed on the edge of the discs and are in phase due to the mirror
symmetry between the two elements. As a consequence, in this case the differential antenna
behaves as a broad-side array, as shown by the radiation pattern in Fig. 23(b). Viceversa,
when the two input ports are fed in phase the currents on the two microstrips are obviously
in phase, whereas on the radiating elements they are out of phase. Indeed, the radiation
pattern in Fig. 24(b) exhibits two radiation lobes and no enhancement of directivity, as
predicted by the calculation of the array factor. Notice that the increase of gain in the case of
differential input with respect to excitation of a single port is about 3 dB, which is exactly
the value predicted by the theory of antenna arrays for a two-element array.
